While most electrical machines operate with alternating currents in their windings, some types of machine are operated with unidirectional current. These include DC machines and doubly salient reluctance machines. In general, a reluctance machine is an electrical machine in which torque is produced by the tendency of its movable part to move into a position where the reluctance of a magnetic circuit is minimised, i.e. where the inductance of the exciting winding is maximised. In some forms, circuitry is provided for detecting the angular position of the rotor and energising the phase windings as a function of the rotor position. This type of reluctance machine is generally known as a switched reluctance machine and it may be operated as a motor or a generator. A general treatment of electrical drives which incorporate switched reluctance machines can be found in various textbooks, e.g. “Electronic Control of Switched Reluctance Machines” by THE Miller, Newnes, 2001, incorporated herein by reference. The characteristics of such switched reluctance machines are well known and are described in, for example, “The Characteristics, Design and Application of Switched Reluctance Motors and Drives” by Stephenson and Blake, PCIM'93, Nürnberg, 21-24 Jun. 1993, incorporated herein by reference. That paper describes in some detail the features of the switched reluctance machine which together produce the characteristic cyclically varying inductance of the phase windings. It is well-known in the art that such machines can be operated in either the motoring or generating mode simply by altering the timing of the winding excitation.
FIG. 1 shows the principal components of a typical switched reluctance drive system, connected to a load 19. The input DC power supply 11 is typically derived from a battery or rectified and filtered AC mains and may be of fixed or variable voltage. The DC voltage provided by the power supply 11 is switched across the phase windings 16 of the motor 12 by a power converter 13 under the control of the electronic control unit 14. The switching must be correctly synchronised to the angle of rotation of the rotor for proper operation of the drive. A rotor position detector 15 is traditionally employed to supply signals indicating the angular position of the rotor. The output of the rotor position detector 15 may also be used to generate a speed feedback signal.
The energisation of the phase windings in a switched reluctance machine depends on detection of the angular position of the rotor. This may be explained by reference to FIGS. 2 and 3, which illustrate the switching of a reluctance machine operating as a motor. FIG. 2 generally shows a rotor pole 20 approaching a stator pole 21 generally in the direction indicated by arrow 22. As illustrated in FIG. 2, a portion 23 of a complete phase winding 16 is wound around the stator pole 21. When the portion 23 of the phase winding 16 around stator pole 21 is energised, a force will be exerted on the rotor, tending to pull rotor pole 20 into alignment with stator pole 21. FIG. 3 generally shows typical switching circuitry in the power converter 13 that controls the energisation of the phase winding 16, including the portion 23 around stator pole 21. The voltage busses 36, 37 are generally known as the DC link and the capacitor 35 across them is known as the DC link capacitor, whose function is to handle the alternating currents on the DC link. When switches 31 and 32 are closed, the phase winding is coupled to the source of DC power and is energised. When the phase winding of a switched reluctance machine is energised in the manner described above, the magnetic field set up by the flux in the magnetic circuit gives rise to the circumferential forces which, as described, act to pull the rotor poles into line with the stator poles.
In general, the phase winding is energised to effect the rotation of the rotor as follows. At a first angular position of the rotor (called the “turn-on angle”, θon), the controller 14 provides switching signals to turn on both switching devices 31 and 32. When the switching devices 31 and 32 are on, the phase winding is coupled to the DC link, causing an increasing magnetic flux to be established in the machine. The magnetic flux produces a magnetic field in the air gap which acts on the rotor poles to produce the motoring torque. The magnetic flux in the machine is supported by the magneto-motive force (mmf) which is provided by a current flowing from the DC supply 11 through the switches 31 and 32 and the phase winding 23. Current feedback is generally employed and the magnitude of the phase current is controlled by chopping the current by rapidly opening or closing one or both of switching devices 31 and/or 32. FIG. 4(a) shows a typical current waveform in the chopping mode of operation, where the current is chopped between two fixed levels. In motoring operation, the turn-on angle θon is often chosen to be the rotor position where the centre-line of an inter-polar space on the rotor is aligned with the centre-line of a stator pole, but may be some other angle. FIG. 4(a) also shows the form of the idealised inductance profile of the phase winding.
In many systems, the phase winding remains connected to the DC link (or connected intermittently if chopping is employed) until the rotor rotates such that it reaches what is referred to as the “freewheeling angle”, θfw. When the rotor reaches an angular position corresponding to the freewheeling angle (e.g., the position shown in FIG. 2) one of the switches, for example 31, is turned off. Consequently, the current flowing through the phase winding will continue to flow, but will now flow through only one of the switches (in this example 32) and through only one of the diodes 33/34 (in this example 34). During the freewheeling period, the voltage drop across the phase winding is small, and the flux remains substantially constant. The circuit remains in this freewheeling condition until the rotor rotates to an angular position known as the “turn-off angle”, θoff, (e.g. when the centre-line of the rotor pole is aligned with that of the stator pole). When the rotor reaches the turn-off angle, both switches 31 and 32 are turned off and the current in phase winding 23 begins to flow through diodes 33 and 34. The diodes 33 and 34 then apply the DC voltage from the DC link in the opposite sense, causing the magnetic flux in the machine (and therefore the phase current) to decrease.
It is known in the art to use other switching angles and other current control regimes. Similarly, many other configurations of lamination geometry, winding topology and switching circuitry are known in the art, some of which are discussed in the Stephenson & Blake paper cited above.
As the speed of the machine rises, there is less time for the current to rise to the chopping level, and the drive is normally run in a “single-pulse” mode of operation. In this mode, the turn-on, freewheel and turn-off angles are chosen as a function of, for example, speed and load torque. Some systems do not use an angular period of freewheeling, i.e. switches 31 and 32 are switched on and off simultaneously. FIG. 4(b) shows a typical such single-pulse current waveform where the freewheel angle is zero. It is well known that the values of turn-on, freewheel and turn-off angles can be predetermined and stored in some suitable format for retrieval by the control system as required, or can be calculated or deduced in real time.
It will be noted that, in both the chopping and single-pulse modes of operation, the current in the phase winding is unidirectional. Mathematically, this can be represented by a zero-frequency component (so-called “DC component” or “mean value”) and series of components at higher frequencies. This is an important distinction from other electrical machines where there is no DC component present.
Strictly speaking, the reference to a zero-frequency component above assumes steady-state operation at constant speed or output and winding temperature (with the average current drawn being constant). When operating conditions are not constant, the “zero-frequency” component will in effect be a low-frequency component with a frequency content determined at least in part by the time constant of changes in operating conditions. In any case, the frequency content of the “zero-frequency” or low-frequency component will be at significantly lower frequencies than the high-frequency components referred to above, which are due to factors such as the switching sequence of actuating the switches to energise the phase winding in question, as well as noise and other high frequency perturbation.
The phase current thus has a low-frequency component at least in part due to changing operating conditions, which has a frequency content below a notional limit frequency. In stable, steady state operation (eg constant speed, output and temperature), the low-frequency component is a substantially zero-frequency component, constant or time-invariant component. The phase current also has a high-frequency component at least in part due to switch actuation, above the notional limit frequency. For convenience of exposition, the terms “zero-frequency component”, “DC component”, “mean component”, “mean value”, “low-frequency component”, “zero-frequency component”, “constant component”, “time invariant component” etc. are used interchangeably in what follows.
In the operation and control of electrical drive systems, a knowledge of the phase winding resistance is often desirable, if not essential. For example, many such systems incorporate methods of estimating the rotor position and many of these methods rely on an accurate measurement of resistance. In other systems, a limit is placed on the temperature at which the windings operate, so as to maintain an acceptable lifetime for the insulation system.
Since the windings are typically based on copper, which has a known temperature coefficient of resistance of around 0.0039, or aluminium, which has a known temperature coefficient of resistance of around 0.0043, it is possible to calculate the average temperature of the phase winding by measuring the resistance at a known temperature (typically an ambient temperature of 20° C.) and measuring or estimating the resistance at the elevated temperature. The procedure for doing this is incorporated in many standards and formal test methods, so that a consistent method of estimating temperature is obtained. For example, Section 8 of IEC 60034-1, “Rotating electrical machines—Part 1: Rating and Performance”, is devoted to the determination of the thermal performance of machines and suggests the use of Equation 1 to determine the average winding temperature rise above ambient:θ2−θa=(R2−R1)/R1*(k+θ1)+θ1−θa  (1)where    θ1 is the temperature (° C.) of the winding (cold) at the moment of the initial resistance measurement;    θ2 is the temperature (° C.) of the winding at the end of the thermal test;    θa is the temperature (° C.) of the coolant at the end of the thermal test;    R1 is the resistance of the winding at temperature θ1 (cold);    R2 is the resistance of the winding at the end of the thermal test;    k is the reciprocal of the temperature coefficient of resistance at 0° C. of the conductor material. (For copper k=235, for aluminium k=225.)
For small machines where the phase winding resistance is in the range, say, 10 to 100Ω, the resistance can be measured by multimeter-type laboratory instruments, whereas for larger machines with correspondingly lower resistances, the use of a four-terminal bridge (e.g. a Kelvin bridge) is generally required to give the required degree of accuracy.
This well-known “rise by resistance” technique for estimating winding temperatures in conventional machines (such as induction motors) requires that the machines be de-energised before resistance measurements are made. Furthermore, the inevitable presence of some electromagnetic saliency within the machine (even if only small, e.g. parasitic effects due to rotor bar geometry and slotting) means that the rotor must generally be brought to a complete standstill before dependable resistance measurements can be made. The traditional technique therefore involves shutting down the drive system, bringing the rotor to rest, plotting a curve of resistance versus time, extrapolating that curve back to the moment of switch-off, and then finally calculating a temperature rise based on the extrapolated value of resistance. Although it is the de facto standard method for gauging the thermal rating of industrial motors, this technique is not only cumbersome and error prone, but cannot be applied to the machine on an on-going basis during normal operation. Other sensing means (such as thermocouples, thermistors, etc.) are additionally required for monitoring winding temperatures whilst the machine is turning and/or energised. Such temperature sensors are often inaccurate (depending as they do on ill-defined thermal contact with the electrically-insulated windings) and may exhibit some time lag compared with the actual or true winding temperature. Furthermore, they require additional low-voltage (and therefore potentially fragile and sensitive) cabling and their additional cost may well be significant in the context of a mass-produced design. There is therefore a need for a non-intrusive and inexpensive means of obtaining an accurate measurement (or estimate) of winding temperature whilst the machine is turning and/or is energised.
Methods of compensating for change in resistance in the windings of other types of electrical machines are known. For example, U.S. Pat. No. 4,496,898 discloses a method of compensation for the temperature rise in the field winding of an ac generator. Methods of rotor resistance compensation in induction machines are known and commonly applied in so-called vector control systems. None of these schemes, however, are applicable to switched reluctance systems, since the effect of the change in resistance is unique to this genre of machine.
There is thus a need for a reliable and economic method of estimating the phase winding resistance of an electrical machine which can operate over all conditions of load (including transient load disturbances) and a wide range of ambient temperatures. The present disclosure is generally applicable to switched reluctance machines operating as motors or generators.